Pulse width modulated power inverter output control

ABSTRACT

A method is provided for output control of a pulse width modulated power inverter used with a substantially resistive, single phase, ac load such as an electro-slag remelting furnace. In one application of the method, during each of the inverter&#39;s half cycle outputs, active switching devices are alternatively pulsed on and off during the half cycle so that only half of the switching transients in the half cycle are handled by one of the active switching devices.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application No.60/711,348 filed Aug. 25, 2005, hereby incorporated herein by referencein its entirety.

FIELD OF THE INVENTION

The present invention relates to the control of the output of a pulsewidth modulated (PWM) power inverter, and in particular, to a powerinverter used with an electro-slag remelting furnace.

BACKGROUND OF THE INVENTION

Electro-slag remelting furnaces require a single phase, ac supply withprecise regulation of load current or power. See U.S. Pat. No. 4,280,550for a description of an electro-slag remelting furnace. In the typicalarrangement of an electro-slag remelting furnace, an electrode isvertically arranged above, but in contact with, a quantity of liquidslag inside a water cooled copper crucible. Power is connected to thetop end of the electrode and the bottom of the crucible such that thecurrent flows through the electrode and slag to the crucible. In thisarrangement the slag acts essentially as a resistance heating element.The heat generated by the current flowing through the slag tends to meltoff the tip of the electrode in contact with the slag and the dropletsof liquid metal, being heavier than the slag, pass through the slag toform a metal pool under the slag which later solidifies into an ingot.In this manner the electrode is progressively melted (remelted) to bereformed as the ingot under the slag. Contact between the liquid metaland the liquid slag tends to refine the metal by removing inclusionssuch that the metallurgical quality of the ingot formed is superior tothat of the electrode remelted, however, precise regulation of furnacecurrent or power is essential to control of the metallurgical quality ofthe ingot formed in the electro-slag remelting furnace.

Traditional methods of controlling the power to an electro-slagremelting furnace includes transformer tap changing, but, this methodrequires frequent maintenance and does not typically achieve therequired precision. The more common method of regulating the current isby phase angle control, but, this method tends to generate high levelsof harmonics to the supply. The combination of a single phase load withhigh harmonics in the approximate range of 800 kilo-watts through 6,000kilo-watts tends to be problematic for modern three phase electricalsupply grids. Consequently there is a need for a better method ofsupplying and regulating the single phase current to an electro-slagremelting furnace.

One method of supplying and regulating the single phase current or powerto an electro-slag remelting furnace is via a PWM inverter. FIG. 1 showsthe output stage of a conventional PWM inverter having a full outputbridge (also known as an “H-bridge”). Energy is stored as a dc voltageon the capacitor C. The switching devices (in this non-limiting exampleshown as IGBT devices) are typically switched on in pairs such thatswitching devices 1 and 4 are switched on for currents of positivepolarity as illustrated by the arrows representing direction of currentflow in FIG. 2( a), and switching devices 2 and 3 are switched on forcurrents of negative polarity as represented in FIG. 2( b). In each casethe switching is controlled so as to achieve an output waveform of thedesired frequency and shape, typically a sine wave. The switchingcontrol may typically be set to output a fixed number of pulses of givenduration in each positive and negative half cycle, respectively, tocontrol the output frequency. The width of the on portion of each pulseis then used to dynamically control both the output waveform andamplitude, respectively.

The electric load of an electro-slag remelting furnace is largelyresistive (resistance R in the figures), but will also include aninductive component (inductance L in the figures). One of the effects ofthe inductance is that when the respective switching devices are turnedoff, the inductance tends to try to maintain the current flow. Ifappropriate measures were not taken this would generate dangerously highvoltages, which could destroy the switching devices. To avoid this, adiode D1, D2, D3 and D4, is connected anti-parallel across each of theswitching devices 1, 2, 3 and 4, respectively, as shown in the figures.In the normal way, the diode polarity is arranged such that the voltageon the capacitor C does not cause current to flow through the diodes.But when the active switches are turned off (not conducting), thereverse voltage generated by the inductance causes two of the diodes tobecome forward biased, and thus conduct the energy stored in theinductive element (inductor) back to the main storage capacitor C. SeeFIG. 3, for example, when switching devices 1 and 4 are just turned off.This effectively limits the voltage generated and thus avoids thedestruction of the switching devices.

The problem is that when switching between a positive and a negativevoltage or current, it is advantageous to discharge the energy stored inthe inductance of the load back into the main storage capacitor, butwhen pulsing one side (polarity) it may, in some designs, become alimiting factor applied to the selection of the diode and the maximumfrequency of switching of the inverter (and hence the precision ofcontrol).

A method is known whereby when the polarity is changed, both activeswitching devices turn off and are made inactive, and then the other twoswitching devices are made active. But, while switching in the givenactive polarity, one of the active switching devices is left on and theother is switched on and off according to the PWM. An example is shownin FIG. 4 where the case is considered that in the particular polarity,the active switching devices are switching devices 1 and 4, andswitching device 1 is left on while the PWM control signal is applied toswitching device 4. In this case when switching device 4 is turned offand switching device 1 remains on, the diode (D3) associated withswitching device 3 acts as a free wheeling diode and the conventionalcurrent flow is as shown in FIG. 4.

Because switching device 1 has a relatively low on (conducting)resistance and the forward voltage of a single diode is relatively low,in this mode the energy stored in the inductor L is dischargedrelatively slowly. This means that current fluctuation due to PWM isreduced.

The disadvantage of this method is that, of the two active switchingdevices, one handles all the switching transients. In designs where theswitching transients are a very large part of the thermal load on theswitching devices, such as in high power inverters, this can cause anuneven temperature distribution in the respective switching devices andpossibly early failure.

There exists, therefore, a need for a method for controlling an inverterused, for example, in an application such as a power supply for anelectro-slag remelting furnace whereby the above limitation is minimizedand the switching transient loads are more evenly distributed.

BRIEF SUMMARY OF THE INVENTION

In one aspect, the present invention is a method of controlling theoutput of a pulse width modulated inverter used, for example, as a powersupply for an electro-slag remelting furnace. The inverter is formedfrom an H-bridge circuit that has at least one switching element inanti-parallel with a diode in each leg of the H-bridge circuit. Asubstantially resistive, single phase load is connected between the legsof the H-bridge. A first pair of switching elements in the first andsecond legs of the H-bridge alternatively conduct for the positive halfcycle of the output, and a second pair of switching elements in thethird and fourth legs of the H-bridge alternatively conduct for thenegative half cycle of the output to establish a flow of ac currentthrough the load. One of the first pair of switching elements is pulsedoff and on for at least one first pulse period of the positive halfcycle while the other one of the first pair of switching elements isconducting, but, then the second switching element of the first pair ispulsed off and on for at least one second pulse period of the positivehalf cycle while the first switching element of the first pair is stillconducting. This alternative switching off and on of one of the twoswitching elements of the first pair while the other is conducting isrepeated for the defined number of pulses of the positive half cycle ofthe output. The first pair of switching elements are then turned off andthe second pair of switching elements is turned on for the negative halfcycle of the output. The second pair of switching elements is controlledin a similar manner by alternatively pulsing off and on one of theswitching elements while the other is conducting until the definednumber of pulses for the negative half cycle is completed. The totalnumber of pulses in both positive and negative half cycles, each of agiven duration, is used to control the output frequency. By dynamicallycontrolling the ratio of on time to off time of each pulse the outputwaveform and amplitude is controlled, but because the switching elementsswitch off and on alternatively the number of switching transients ineach switching element is reduced.

Other aspects of the invention are set forth in this specification andappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For the purpose of illustrating the invention, there is shown in thedrawings a form that is presently preferred; it being understood,however, that this invention is not limited to the precise arrangementsand instrumentalities shown.

FIG. 1 is a schematic of the output stage of a typical power inverter.

FIG. 2( a) and FIG. 2( b) illustrate direction of current flow in theoutput stage of the inverter shown in FIG. 1 with switching devices 1and 4 on, and switching devices 3 and 2 on, respectively.

FIG. 3 illustrates current flow in the output stage of the invertershown in FIG. 1 when switching devices 1 and 4 are just turned off.

FIG. 4 illustrates current flow in the output stage of the invertershown in FIG. 1 with switching device 1 conducting and switching device4 just turned off, after switching devices 1 and 4 were on.

FIG. 5 illustrates current flow in the output stage of the invertershown in FIG. 1 with switching device 4 conducting and switching device1 just turned off, after switching devices 1 and 4 were on.

FIG. 6 illustrates the schematic for a typical single phase, full “H”bridge ac inverter used for the control scheme of the present invention.

FIG. 7 illustrates in tabular form the bi-phase PWM control of oneexample of the present invention in terms of the logic levels applied tothe gates of respective switching devices.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 6 illustrates one application in which the PWM inverter outputcontrol method of the present invention can be used. DC power isinputted to inverter 6 from a single phase rectifier formed from diodesD5, D6, D7 and D8, with capacitor C serving as an energy storage andfilter element. PWM control circuit 8 includes gate driver circuitry forgates G1, G2, G3 and G4 of switching devices 1, 2, 3 and 4,respectively. Although IGBT switching devices are illustrated in FIG. 6,other types of switching devices may be used. Each of the switchingdevices has an allowed switching rate, as defined by the specificationfor a specific switching device. The gate driver circuitry controls turnon and turn off of the switching devices, including alternative pulsingof the switching devices as further described below. In some examples ofthe invention, suitable circuitry provides inputs to the PWM controlcircuit 8 for regulating the alternative pulsing scheme, including pulsefrequency and pulse duration. For example the magnitude of outputcurrent or power from the inverter, voltage across the output of theinverter, or voltage across the capacitor, may be inputted to the PWMcontrol circuit. In some examples of the invention the disclosed pulsingscheme may be dynamically adjusted according to the instantaneous valueof the inverter's output current or voltage. Further the dynamicadjustment may be accomplished according to the amplitude of the RMSvalue of the inverter's output current or voltage, which can bedetermined, for example, from the instantaneous value of the inverter'soutput current or voltage, respectively.

In FIG. 6, one type of load, namely an electro-slag remelting furnace,that can be used with the PWM inverter output control method of thepresent invention is illustrated. Optional transformer, T, provides acpower from the output of inverter 6 at circuit nodes “A” and “B”. Thetransformer, if used, may be any type of kVA transformer element, orcombination of elements. The load, electro-slag remelting furnace 10, issimilar to the furnace disclosed in U.S. Pat. No. 4,280,550 (the 550patent), which is hereby incorporated herein by reference in itsentirety. Element numbers for the following features of the furnace arethe same as those used in the 550 patent. Electrode 32 of the furnace isconnected to conductor 52 as further described in the 550 patent, andconductor 52, in turn, is connected to one output connection of theinverter. Baseplate 38 is connected by conductor 50 to the other outputconnection of the inverter.

In other examples of the invention, a direct dc input to the invertermay be provided. Where the input is ac, the input may be supplied from asingle phase source, as shown in FIG. 6, or a multi-phase ac source. Therectifier element may be of any suitable form when an ac input is used.

The output stage of inverter 6 comprises a number of switching devicesthat are respectively controlled to create an inverter ac output of adesired frequency, amplitude and waveform. Although one H-bridgearrangement is illustrated in FIG. 6, in other examples of theinvention, multiple H-bridges, suitably arranged, may be used.

The load connected to the inverter is a static (non-rotating) largelyresistive load, which, for example, is typical of the loading in anelectro-slag remelting furnace. The inductive load component of theelectro-slag remelting furnace tends to be reduced by the coaxialarrangement of furnace elements, but will vary with furnace size andelectrode diameter, and so forth. Reference to inductance, or inductorL, relative to FIG. 6, refers to this inductive load component for thisparticular example of the invention. Inverter design and control must beable to handle this complex load and must also be able to operate duringthe startup phase of the process (cold start or hot start, respectively)and the hot topping or ramp down phase at the end of the process. Coldstarting of the electro-slag remelting furnace, in particular, mayinvolve periods of arcing and sudden fluctuation of the load from shortcircuit to open circuit, and visa versa, as the slag is initiallymelted. Even during the so called steady state portion of the remelt inan electro-slag remelting furnace, fluctuation of the contact areabetween the liquid tip of the electrode and the liquid slag causes aconstantly fluctuating load impedance requiring dynamic control of loadcurrent. Because an electro-slag remelting furnace will typicallyrequire a supply rated in the range 800 kilo-watts to 6,000 kilo-watts,careful optimization of control power components is essential toreliable operation.

In the present invention, a bi-phase PWM output control of the inverteris utilized wherein the two active switching devices, or elements, arealternatively pulsed off while the other active switching device is heldon, and visa versa.

In one non-limiting example of the invention, switching devices 1 and 4are the active switching devices during the present half cycle of theoutput of the inverter, which is defined as the positive half cycle forthe non-limiting circuit arrangement in FIG. 6. During the positive halfcycle, while switching device 1 is on, switching device 4 is pulsed offfor a period of “device 4 off time” and then turned on; then whileswitching device 4 is on, switching device 1 is pulsed off for a period“device 1 off time.” This process of alternatively pulsing switchingdevice 4 and switching device 1 off during the positive half cycle maybe sequentially repeated throughout the positive half cycle.

In other examples of the invention, switching device 4 may be repeatedlypulsed off and on multiple times before switching device 1 is pulsed offand then on, at least once during the positive half cycle.

In some examples of the invention, the period of switching device 4 offtime may be different from the period of switching device 1 off timeand/or the number of switching device 4 off time pulses may be differentfrom the number of switching device 1 off time pulses.

With the control method of the present invention, each time switchingdevice 4 is pulsed off during the positive half cycle while switchingdevice 1 is on, the switching transients are handled by switching device4 while the circuit point of the load labeled “A” is effectively heldclose to the positive bus (+BUS) voltage by switching device 1, and theback electromagnetic force generated by the energy stored in theinductor L forward biases the anti-parallel diode D3 of switching device3, which clamps load circuit point “B” near to the same positive busvoltage. Thus the energy stored in inductance L is discharged at arelatively low voltage which depends on the ratio of diode voltage tobus voltage. Conversely, in the alternate phase of the bi-phase PWMcontrol of the present invention, each time switching device 1 is pulsedoff during the positive half cycle while switching device 4 is on, theswitching transients are handled by switching device 1 while the circuitpoint of the load labeled “B” is effectively held close to the negativebus (−BUS) voltage by switching device 4, and the back electromagneticforce generated by the energy stored in the inductor L forward biasesthe anti-parallel diode D2 of switching device 2, which clamps loadcircuit point “B” near to the same negative bus voltage. Thus the energystored in inductance L is similarly discharged at a relatively lowvoltage.

The above description applies to the positive half cycle in thisnon-limiting example of the invention, when switching devices 1 and 4are the active switching devices. In the same example, switching devices3 and 2 are the active switching devices during the negative half cycle.During the negative half cycle, while switching device 3 is on,switching device 2 is pulsed off for a period of “device 2 off time” andthen turned on; then while switching device 2 is on, switching device 3is pulsed off for a period of “device 3 off time.” This process ofalternatively pulsing switching device 2 and switching device 3 offduring the negative half cycle may be sequentially repeated throughoutthe negative half cycle.

In other examples of the invention, switching device 2 may be repeatedlypulsed off and on multiple times before switching device 3 is pulsed offand then on, at least once during the positive half cycle.

In some examples of the invention the period of switching device 2 offtime may be different from the period of switching device 3 off timeand/or the number of switching device 2 off time pulses may be differentfrom the number of switching device 3 off time pulses.

During the negative half cycle, each time switching device 2 is pulsedoff while switching device 3 is on, the switching transients are handledby switching device 2 while the circuit point of the load labeled “B” iseffectively held close to the positive bus voltage by switching device3, and the back electromagnetic force generated by the energy stored inthe inductor L forward biases the anti-parallel diode D1 of switchingdevice 1, which clamps load circuit point “A” near to the same positivebus voltage. Thus the energy stored in inductance L is discharged at arelatively low voltage which depends on the ratio of diode voltage tobus voltage. Conversely, in the alternate phase of the bi-phase PWMcontrol of the present invention, each time switching device 3 is pulsedoff during the negative half cycle while switching device 2 is on, theswitching transients are handled by switching device 3 while the circuitpoint of the load labeled “A” is effectively held close to the negativebus voltage by switching device 2, and the back electromagnetic forcegenerated by the energy stored in the inductor L forward biases theanti-parallel diode D4 of switching device 4, which clamps load circuitpoint “B” near to the same negative bus voltage. Thus the energy storedin inductance L is similarly discharged at a relatively low voltage.

The significant advantage is that with bi-phase PWM inverter outputcontrol of the present invention, effectively only half of the switchingtransients in a given half cycle are handled by one of the activeswitching devices. For high power inverters handling a load in thehundreds or thousands of kilo-watts, the switching transients are one ofthe design factors limiting a given design using specific devices. Thusfor a given set of switching devices the benefits of the presentinvention include: the thermal heat load on each device may be reduce;or the effective reliability of the inverter may be improved; or theload rating may be increase; or the effective switching rate of the PWMsignal may be increased; or a combination of any of the above.

In the above examples of the invention, the term “PWM” is used todescribe the general type or class of inverter but is not intended tolimit the present invention to the specific class of inverter whereinthe time interval of successive pulses is controlled to generate aspecific output waveform, for example a sine wave. The present inventionincludes, but not be limited to, those forms of PWM where the pulsewidth or switching point is determined by comparing the actual (ormeasured) output voltage or current to a reference voltage or current(digital or analog).

FIG. 7 illustrates in tabular form the bi-phase PWM inverter outputcontrol of one example of the present invention in terms of the logiclevels applied to the respective switch device gates. A standard pulsewidth (labeled “NOMINAL PWM (COMPOSITE)”) is shown for clarity and toindicate that the present invention is not specifically limited to aparticular output wave form or method of deriving the PWM signal, but israther concerned with the method of applying the signal to respectiveactive devices in an alternating sequence which achieves the benefitsindicated. The gate signals of the present invention for switchingdevice 1 (column “BI-PHASE PWM-1”); switching device 2 (column “BI-PHASEPWM-2”); switching device 3 (column “BI-PHASE PWM-3”); and switchingdevice 4 (column “BI-PHASE PWM-4”) illustrated in FIG. 7 can be providedto the respective switching devices by circuitry known in the art.

The present invention will be most efficacious for large high powerinverters in the range from approximately 800 kilo-watts toapproximately 6,000 kilo-watts typically used with static (non-rotating)largely resistive single phase ac loads wherein the switching transientswill be effectively reduced in each device, which will achieve higherreliability or higher effective PWM switching rates.

The examples of the invention include reference to specific electricalcomponents. One skilled in the art may practice the invention bysubstituting components that are not necessarily of the same type butwill create the desired conditions or accomplish the desired results ofthe invention. For example, single components may be substituted formultiple components or vice versa. Further one skilled in the art maypractice the invention by rearranging components to create the desiredconditions or accomplish the desired results of the invention.

The foregoing examples do not limit the scope of the disclosedinvention. The scope of the disclosed invention is further set forth inthe appended claims.

1. A method of controlling the output of a pulse width modulatedinverter comprising at least one H-bridge circuit having a first andsecond switching elements in a first leg of the H-bridge circuit and athird and fourth switching elements in a second leg of the H-bridgecircuit, an anti-parallel diode connected across each one of the first,second, third and fourth switching elements, the first and thirdswitching elements connected to a positive bus of the pulse widthmodulated inverter and the second and fourth switching elementsconnected to a negative bus of the pulse width modulated inverter, afirst load terminal at a common connection between the first and secondswitching elements and a second load terminal at a common connectionbetween the third and fourth switching elements, each of the first,second, third and fourth switching elements having an allowed switchingrate, the output of the inverter connected to a load connected betweenthe first and second load terminals, the load comprising a substantiallyresistive, single phase load, the method comprising the steps ofcontrolling current to the load by the first and fourth switchingelements during the positive half cycle of the output waveform, andcontrolling current to the load by the second and third switchingelements during the negative half cycle of the output waveform, toestablish the flow of an ac current through the load, an improvementcomprising, turning on the first and fourth switching elements duringthe positive half cycle while the second and third switching elementsare turned off and alternatively pulsing the first switching element offand on for at least one first pulse period, at least once during thepositive half cycle while the fourth switching element is turned on, andalternatively pulsing the fourth switching element off and on for atleast one second pulse period, at least once during the positive halfcycle while the first switching element is turned on, and turning on thesecond and third switching elements during the negative half cycle whilethe first and fourth switching elements are turned off and alternativelypulsing the third switching element off and on for at least one thirdpulse period, at least once during the negative half cycle while thesecond switching element is turned on, and alternatively pulsing thesecond switching element off and on for at least one fourth pulseperiod, at least once during the negative half cycle while the thirdswitching element is turned on.
 2. The method of claim 1 wherein thesteps of alternatively pulsing the first and fourth switching elements,and alternatively pulsing the second and third switching elements arearranged to reduce the number of switching transients in the first andfourth switching elements or the second and third switching elements,respectively.
 3. The method of claim 1 further comprising the step ofadjusting the number of the first, second, third and fourth pulseperiods responsive to the allowed switching rate.
 4. The method of claim3 further comprising the step of adjusting the duration of the first,second, third and fourth pulse periods to control the frequency waveformof the output of the inverter.
 5. The method of claim 4 furthercomprising the step of dynamically adjusting the ratio of the on time tooff time of each of the first, second, third and fourth pulse periods tocontrol the current waveform of the output of the inverter toapproximate that of a sine wave.
 6. The method of claim 5 furthercomprising the step of adjusting the ratio of the on time to the offtime of each of the first, second, third and fourth pulse periods tocontrol the amplitude of the RMS value of current output of theinverter.
 7. The method of claim 6 further comprising the step ofoperating the inverter with output power in the range from approximately800 kilo-watts to approximately 6,000 kilo-watts.
 8. The method of claim4 further comprising the step of dynamically adjusting the ratio of theon time to off time of each of the first, second, third and fourth pulseperiods to control the voltage waveform of the output of the inverter toapproximate that of a sine wave.
 9. The method of claim 8 furthercomprising the step of adjusting the ratio of the on time to the offtime of each of the first, second, third and fourth pulse periods tocontrol the amplitude of the RMS value of voltage output of theinverter.
 10. The method of claim 9 further comprising the step ofoperating the inverter with output power approximately in the range fromapproximately 800 kilo-watts to approximately 6,000 kilo-watts.
 11. Amethod of controlling current or power to an electro-slag remeltingfurnace from a pulse width modulated inverter comprising at least oneH-bridge circuit having a first and second switching elements in a firstleg of the H-bridge circuit and a third and fourth switching elements ina second leg of the H-bridge circuit, an anti-parallel diode connectedacross each one of the first, second, third and fourth switchingelements, the first and third switching elements connected to a positivebus of the pulse width modulated inverter and the second and fourthswitching elements connected to a negative bus of the pulse widthmodulated inverter, a first load terminal at a common connection betweenthe first and second switching elements and a second load terminal at acommon connection between the third and fourth switching elements, eachof the first, second, third and fourth switching elements having anallowed switching rate, the output of the inverter connected to a loadconnected between the first and second load terminals, the loadcomprising a substantially resistive, single phase load, the methodcomprising the steps of controlling current to the load by the first andfourth switching elements during the positive half cycle of the outputwaveform, and controlling current to the load by the second and thirdswitching elements during the negative half cycle of the outputwaveform, to establish the flow of an ac current through the load, animprovement comprising, turning on the first and fourth switchingelements during the positive half cycle while the second and thirdswitching elements are turned off and alternatively pulsing the firstswitching element off and on for at least one first pulse period, atleast once during the positive half cycle while the fourth switchingelement is turned on, and alternatively pulsing the fourth switchingelement off and on for at least one second pulse period, at least onceduring the positive half cycle while the first switching element isturned on, and turning on the second and third switching elements duringthe negative half cycle while the first and fourth switching elementsare turned off and alternatively pulsing the third switching element offand on for at least one third pulse period, at least once during thenegative half cycle while the second switching element is turned on, andalternatively pulsing the second switching element off and on for atleast one fourth pulse period, at least once during the negative halfcycle while the third switching element is turned on.
 12. The method ofclaim 11 wherein the steps of alternatively pulsing the first and fourthswitching elements, and alternatively pulsing the second and thirdswitching elements are arranged to reduce the number of switchingtransients in the first and fourth switching elements or the second andthird switching elements, respectively.
 13. The method of claim 11further comprising the step of adjusting the number of the first,second, third and fourth pulse periods responsive to the allowedswitching rate.
 14. The method of claim 13 further comprising the stepof adjusting the duration of the first, second, third and fourth pulseperiods to control the frequency waveform of the output of the inverter.15. The method of claim 14 further comprising the step of dynamicallyadjusting the ratio of the on time to off time of each of the first,second, third and fourth pulse periods to control the current waveformof the output of the inverter to approximate that of a sine wave. 16.The method of claim 15 further comprising the step of adjusting theratio of the on time to the off time of each of the first, second, thirdand fourth pulse periods to control the amplitude of the RMS value ofcurrent output of the inverter.
 17. The method of claim 16 furthercomprising the step of operating the inverter with output power in therange from approximately 800 kilo-watts to approximately 6,000kilo-watts.
 18. The method of claim 14 further comprising the step ofdynamically adjusting the ratio of the on time to off time of each ofthe first, second, third and fourth pulse periods to control the voltagewaveform of the output of the inverter to approximate that of a sinewave.
 19. The method of claim 18 further comprising the step ofadjusting the ratio of the on time to the off time of each of the first,second, third and fourth pulse periods to control the amplitude of theRMS value of voltage output of the inverter.
 20. The method of claim 19further comprising the step of operating the inverter with output powerapproximately in the range from approximately 800 kilo-watts toapproximately 6,000 kilo-watts.
 21. A method of controlling current orpower to an electro-slag remelting furnace from a pulse width modulatedinverter comprising at least one H-bridge circuit having a first andsecond switching elements in a first leg of the H-bridge circuit and athird and fourth switching elements in a second leg of the H-bridgecircuit, an anti-parallel diode connected across each one of the first,second, third and fourth switching elements, the first and thirdswitching elements connected to a positive bus of the pulse widthmodulated inverter and the second and fourth switching elementsconnected to a negative bus of the pulse width modulated inverter, afirst load terminal at a common connection between the first and secondswitching elements and a second load terminal at a common connectionbetween the third and fourth switching elements, each of the first,second, third and fourth switching elements having an allowed switchingrate, the output of the inverter connected to a load connected betweenthe first and second load terminals, the load comprising a substantiallyresistive, single phase load, the method comprising the steps ofcontrolling current to the load by the first and fourth switchingelements during the positive half cycle of the output waveform, andcontrolling current to the load by the second and third switchingelements during the negative half cycle of the output waveform, toestablish the flow of an ac current through the load, an improvementcomprising, discharging load energy during the positive half cycle byalternatively pulsing the first switching element off and on for atleast one first pulse period, at least once during the positive halfcycle while the fourth switching element is turned on, and alternativelypulsing the fourth switching element off and on for at least one secondpulse period, at least once during the positive half cycle while thefirst switching element turned on, and discharging load energy duringthe negative half cycle by alternatively pulsing the third switchingelement off and on for at least one third pulse period, at least onceduring the negative half cycle while the second switching element isturned on, and alternatively pulsing the second switching element offand on for at least one fourth pulse period, at least once during thenegative half cycle while the third switching element is turned on. 22.The method of claim 21 wherein the steps of alternatively pulsing thefirst and fourth switching elements, and alternatively pulsing thesecond and third switching elements are arranged to reduce the number ofswitching transients in the first and fourth switching elements or thesecond and third switching elements, respectively.
 23. The method ofclaim 21 further comprising the step of adjusting the number of thefirst, second, third and fourth pulse periods responsive to the allowedswitching rate.
 24. The method of claim 23 further comprising the stepof adjusting the duration of the first, second, third and fourth pulseperiods to control the frequency waveform of the output of the inverter.25. The method of claim 24 further comprising the step of dynamicallyadjusting the ratio of the on time to off time of each of the first,second, third and fourth pulse periods to control the current waveformof the output of the inverter to approximate that of a sine wave. 26.The method of claim 25 further comprising the step of adjusting theratio of the on time to the off time of each of the first, second, thirdand fourth pulse periods to control the amplitude of the RMS value ofcurrent output of the inverter.
 27. The method of claim 26 furthercomprising the step of operating the inverter with output power in therange from approximately 800 kilo-watts to approximately 6,000kilo-watts.
 28. The method of claim 24 further comprising the step ofdynamically adjusting the ratio of the on time to off time of each ofthe first, second, third and fourth pulse periods to control the voltagewaveform of the output of the inverter to approximate that of a sinewave.
 29. The method of claim 28 further comprising the step ofadjusting the ratio of the on time to the off time of each of the first,second, third and fourth pulse periods to control the amplitude of theRMS value of voltage output of the inverter.
 30. The method of claim 29further comprising the step of operating the inverter with output powerapproximately in the range from approximately 800 kilo-watts toapproximately 6,000 kilo-watts.